Digital phase locked loop

ABSTRACT

A phase locked loop circuit ( 30, 100, 110 ) includes a controllable oscillator ( 42 ) for generating an output signal of desired frequency responsive to a control signal, a first phase detection circuit ( 32, 102, 112 ) for generating an output indicative of phase differential responsive to the output signal and a first edge of a reference signal and a second phase detection circuit ( 34, 104, 114 ) for generating an output indicative of phase differential responsive to the output signal and a second edge of a reference signal. The control signal to the controllable oscillator ( 42 ) is driven by the outputs of the first and second phase detections circuits.

CROSS-REFERENCE TO RELATED APPLICATIONS

This application is a divisional of prior Application No. 14/743,900,filed Jun. 18, 2015, currently pending;

Which was a Divisional of Application No. 14/525,965, filed Oct. 28,2014, now U.S. Pat. No. 9,094,184, granted Jul. 28, 2015;

Which is a divisional of Application No. 13/710,722, filed Dec. 11,2012, now abandoned;

Which is a Continuation of Application No. 10/131,523, filed Apr. 24,2002, now U.S. Pat. No. 8,385,476, granted Feb. 26, 2013;

Which claims the benefit of the filing date of copending provisionalapplication U.S. Ser. No. 60/286,572, filed Apr. 25, 2001, entitled“Frequency Synthesizer Architecture of the Digital Radio Processor(v2.0)” to Staszewski et al.

STATEMENT OF FEDERALLY SPONSORED RESEARCH OR DEVELOPMENT

Not Applicable

BACKGROUND OF THE INVENTION

1. Technical Field

This invention relates in general to electronics and, more particularly,to a digital phase locked loop.

2. Description of the Related Art

A great reduction of the transistor features in recently developeddeep-submicron CMOS processes shifts the design paradigm towards moredigitally-intensive techniques. In a monolithic implementation, themanufacturing cost of a design is measured not in terms of a number ofdevices used but rather in terms of the occupied silicon area, no matterwhat the actual circuit complexity.

Analog and RF circuits used in communication circuits, however, are noteasily implemented in a deep-submicron CMOS process. For example, inTexas Instruments' CMOS process (CO35) of 0.08 □m L-effective features adigital gate density of 150 K equivalent (2-input NAND) gates per mm².An average-size inductor for an integrated LC oscillator occupies about0.5 mm² of silicon area. A low-noise charge pump, or a low-distortionimage-reject modulator, both good examples of classical RF transceivercomponents, occupy roughly about the same area, which could be tradedfor tens of thousands of digital gates.

Migrating to a digitally-intensive synthesizer architecture brings forththe following well-known advantages: (1) fast design turn-around cycleusing automated CAD tools (VHDL or Verilog hardware-level descriptionlanguage, synthesis, auto-place and auto-route with timing-drivenalgorithms, parasitic backannotation and postlayout optimization), (2)much lower parameter variability than with analog circuits, (3) ease oftestability, (4) lower silicon area and dissipated power that getsbetter with each CMOS technology advancement (also called a “processnode”) and (5) excellent chances of first-time silicon success.Commercial analog circuits usually require several design iterations tomeet marketing requirements.

There is a wide array of opportunities that integration presents. Themost straightforward way would be to merge various digital sections intoa single silicon die, such as DRAM or Flash memory embedded into DSP orcontroller. More difficult would be integrating the analog baseband withthe digital baseband. Care must be taken here to avoid coupling ofdigital noise into the high-precision analog section. In addition, thelow amount of voltage headroom challenges one to find new circuit andarchitecture solutions. Integrating the analog baseband into RFtransceiver section presents a different set of challenges: theconventional Bi-CMOS RF process is tuned for high-speed operation with anumber of available passive components and does not fundamentally stresshigh precision.

Sensible integration of diverse sections results in a number ofadvantages: (1) lower total silicon area—in a deep-submicron CMOSdesign, the silicon area is often bond-pad limited; consequently, it isbeneficial to merge various functions on a single silicon die tomaximize the core to bond-pad ratio, (2) lower component count and thuslower packaging cost, (3) power reduction—no need to drive largeexternal inter-chip connections and (4) lower printed-circuit board(PCB) area, thus saving the precious “real estate.”

Deep-submicron CMOS processes present new integration opportunities onone hand, but make it extremely difficult to implement traditionalanalog circuits, on the other. One such problem involves the design of adigital phase locked loop (DPLL). A PLL loop is a fixed-point phasedomain architecture whose purpose is to generate a stable RF signal at adesired frequency. The underlying frequency stability of the system isderived from a reference clock generated by a crystal oscillator, suchas a temperature-compensated crystal oscillator (TCXO) used in mobilephones. Phase information between the output signal and the referencesignal is used to update a controllable oscillator. This information isgenerated at an active edge of the reference clock. However, greateraccuracy could be obtained by more frequent determinations of the phaseinformation, leading to more frequent updates of the controllableoscillator.

Therefore, a need has arisen for a method and apparatus for a phaselocked loop design that provides for increase accuracy in the outputsignal.

BRIEF SUMMARY OF THE INVENTION

In a first aspect of the present invention, a phase locked loop circuitincludes a controllable oscillator for generating an output signal ofdesired frequency, a first phase detection circuit for generating anoutput indicative of phase differential responsive to the output signaland a first edge of a reference signal and a second phase detectioncircuit for generating an output indicative of phase differentialresponsive to the output signal and a second edge of a reference signal.The controllable oscillator is driven responsive to the outputs of thefirst and second phase detections circuits.

This aspect of the invention increases the timing updates for thephase-locked loop since both edges of the reference clock are used forphase detection.

In a second aspect of the present invention, a mobile communicationdevice comprises a frequency synthesizer for generating a carrierfrequency output responsive to a local reference clock, circuitry forgenerating multiple clock signals of different frequencies synchronousto the carrier frequency output and digital baseband circuitry operatingresponsive to one or more of the multiple clock signals.

This aspect of the present invention allows for a plurality of clocks tobe derived from the output of a frequency synthesizer. By reducing thenumber of phase-locked loop circuits use to generate clocks, unnecessarycircuitry can be eliminated. Further, by providing a number of clockssynchronous to the RF carrier frequency, spurious noise throughout themobile communication device can be controlled to reduce the effect oncommunications. An additional benefit is that complex circuitry forsynchronizing the local reference signal to a master clock signal can beeliminated, and the carrier frequency can be synchronized to the masterclock through minor adjustments to a phase error signal.

BRIEF DESCRIPTION OF THE SEVERAL VIEWS OF THE DRAWINGS

For a more complete understanding of the present invention, and theadvantages thereof, reference is now made to the following descriptionstaken in conjunction with the accompanying drawings, in which:

FIG. 1a illustrates a block diagram of a prior art phase locked loopcircuit;

FIG. 1b illustrates a timing diagram showing the operation of thecircuit of FIG. 1 a;

FIG. 2 illustrates a timing diagram showing relationships between therising and falling edges of a reference clock;

FIG. 3a illustrates a block diagram of a phase locked loop circuitcapable of sampling on both edges of a reference clock;

FIG. 3b illustrates a timing diagram showing the operation of the phaselocked loop circuit of FIG. 3 a;

FIG. 4a illustrates a block diagram of an all digital phase locked loop(ADPLL);

FIG. 4b illustrates a timing diagram showing the operation of the ADPLLof FIG. 4 a;

FIG. 5 illustrates an ADPLL using the both edges of the reference clockto double the sampling rate of the PLL operation;

FIG. 6 illustrates a schematic diagram of a time-to-digital converter;

FIG. 7 illustrates a timing diagram showing the operation of thetime-to-digital converter of FIG. 6;

FIG. 8 illustrates a second embodiment of a PLL sampling on both edgesof the reference clock;

FIG. 9 illustrates a third embodiment of a PLL sampling on both edges ofthe reference clock;

FIG. 10 illustrates a general block diagram of a prior art mobilecommunication device;

FIG. 11 illustrates a general block diagram of circuitry for providing aPLL function using a divided clock from the DCO as a synchronous clockfor the RF transceiver and digital baseband circuit;

FIG. 12 illustrates a more detailed block diagram for a first embodimentof the circuit of FIG. 11; and

FIG. 13 illustrates a more detailed block diagram for a secondembodiment of the circuit of FIG. 11.

DETAILED DESCRIPTION OF THE INVENTION

The present invention is best understood in relation to FIGS. 1-13 ofthe drawings, like numerals being used for like elements of the variousdrawings.

FIG. 1a illustrates a block diagram of a generalized phase locked loopdevice (PLL) 10. A reference frequency FREF, typically generated by acrystal oscillator, is input to a phase detector 12 along with CKVD, thedivided-down clock output of the PLL 10. An error signal, Φ, is passedto a loop filter 14. The filtered signal adjusts the output of acontrollable oscillator 16. The output of the controllable oscillator16, CKV, is fed back to the phase detector 12 through a frequencydivider 18.

In general, the phase detector (and, hence, the controllable oscillator)operates responsive to an active edge of the FREF signal. For purposesof illustration throughout this specification, it will be assumed thatthe rising edge of FREF is the active edge; alternatively, the fallingedge could be used as the active edge of FREF.

In many situations, it would be beneficial to compare the phases andupdate the output signal more often. One possible solution would be toclock the phase detector and controllable oscillator at both the risingand falling edges of FREF.

As shown in FIG. 1b , however, clocking on both edges of the referencesignal presents a problem. While the output of a crystal oscillatorproduces a signal with a fairly accurate 50% duty cycle (i.e., the timebetween a rising edge and the subsequent falling edge is exactly thesame as the time between a falling edge and a subsequent rising edge),intervening circuitry can affect the duty cycle, such that the fallingedge may occur within a range 20 in FIG. 1b . Consequently, the fallingedge of FREF cannot be used as a mid-point between rising edges of FREF.

As shown in FIG. 2, the present invention uses a property of FREF togain useful information from the falling edge of FREF. In FIG. 2, dashedline 22 shows the ideal 50% duty cycle mark for the falling edge. Whilethe actual falling edge, shown at line 24 may be offset from the ideal,the time period T_(fall) between actual falling edges 24 equals the timeperiod T_(rise) between rising edges. Thus, for each cycle, the actualfalling edge 24 will be offset from the ideal 22 by a constant ΔΦ.

FIG. 3a illustrates a first embodiment of the invention for using bothedges of a reference clock in a PLL 30. FREF is input to a first phasedetector 32 (which compares phase information on a first active edge,e.g. rising edge) of FREF and a second phase detector 34 (which comparesphase information on a second active edge, e.g. falling edge) of FREF.Alternatively, the FREF signal is input to the first phase detector 32and the inverted FREF signal is input to the second phase detector 34,and both phase detectors operate internally on the same active edge ofthe reference clock signals that are 180 degrees out of phase. Theoutput of the first phase detector 32 is a first error signal Φ₁ and theoutput of the second phase detector 34 is a second intermediate errorsignal Φ₂′. The intermediate error signal Φ₂′ is added to ΔΦ throughphase offset adder 36 to generate the second error signal Φ₂. The firstand second error signals are input to multiplexer 38, which selects oneof the first and second error signals responsive to an rising/falling(R/F) control signal, which indicates whether the current active edge ofFREF is rising or falling. The output of multiplexer 38 is received byloop filter 40. The output of loop filter 40 drives oscillator 42. Theoutput of oscillator 42 (CKV) is received by frequency divider 44. Thedivided output signal CKVD is transmitted to phase detectors 32 and 34.

FIG. 3b illustrates a timing diagram showing FREF, CKVD and two examplesof CKV (for N=2 and N=3). As can be seen in FIG. 3b , when phasedetectors compare FREF and CKVD as shown in FIG. 3a , phase detector 32compares the rising edge of FREF with the rising edge of CKVD and phasedetector 34 compares the falling edge of FREF (or the rising edge of theinverted FREF) with the falling edge of CKVD (or the rising edge of theinverted CKVD). ΔΦ is the phase difference between the actual fallingedge of FREF and the ideal 50% duty cycle falling edge. For a risingedge, Φ_(E), the output of multiplexer 38, equals Φ₁ and, for a fallingedge, Φ_(E) equals Φ₂, which equals Φ₂′+ΔΦ.

It would also be possible to compare the edges of FREF with the outputCKV. In this case, both the rising edge and falling edge of FREF wouldbe compared to a rising edge of CKV, if N (f_(CKV)/f_(FREF)) was an eveninteger. If N is an odd integer, two approaches could be used. In thefirst approach, rising edges of FREF would be compared to rising edgesof CKV and falling edges of FREF would be compared to falling edges ofCKV (it would also be possible to compare rising edges of FREF tofalling edges of CKV and vice-versa). In the second embodiment, ahalf-phase adjustment Φ_(H) could be added to Φ₂′ along with z by thephase offset adder 36, such that Φ₂=Φ₂′+ΔΦ+Φ_(H).

FIG. 4a illustrates a block diagram of an all digital PLL (ADPLL) 60 ofthe type disclosed in U.S. patent application Ser. No. 10/008,462, nowU.S. Pat. No. 7,006,589, to Staszewski et al, entitled “FrequencySynthesizer with Phase Restart”, filed Nov. 30, 2001, which isincorporated by reference herein. This ADPLL is designed to work off asingle edge of FREF.

ADPLL 60 includes a reference phase accumulator 62 that calculates areference phase signal (PHR), a fractional error correction circuit 64that calculates a fractional error correction (PHF), and a variablephase accumulator 66 that calculates a variable phase correction(PHV_SMP, which is integer only). The phase error (PHE) is calculated byphase detector 68 as PHE=PHR+PHF-PHV_SMP (with proper bit alignment toline up integer and fractional portions). PHE is received by gaincircuit 70 and oscillator control circuit 72. The oscillator controlcircuit 72 drives a digitally controlled oscillator 74.

FCW (frequency control word) is the ratio of the desired frequency ofCKV divided by the frequency of FREF. The reference phase signal is anaccumulation of FCW at the active edge of CKR, which is the retimed FREFclock. The FCW input to the reference accumulator 62 is used toestablish the operating reference phase of the desired channel plus themodulation data.

The variable phase accumulator 66 comprises a counter 66 a, whichincrements on each active edge of CKV and a latch 66 b that latches theoutput of the counter at CKR.

The fractional phase circuit 64 determines a difference between anactive edge of FREF and the next active edge of CKV, normalized to afraction of a CKV clock cycle.

Operation of the circuit is best understood in relation to the timingdiagram of FIG. 4b , where an example of FCW=2.25 is used. In an actualcircuit, however, FCW would typically be much higher. For purposes ofillustration, FCW is a constant (i.e., no modulation) and there is nodrift. As described above, the variable phase circuit 66 counts the CKVclocks and latches the count on the active (rising) edge of CKR. The PHVfrom the variable phase circuit 66 counts are provided above the CKVsignal. Also at each active edge of CKR, the reference phase circuit 62accumulates another FCW.

At any active edge of CKR, the preceding active edge of FREF may haveoccurred at a point less than one CKV clock cycle earlier (since CKR isretimed to CKV). This is shown by the dashed lines in FIG. 4b . PHFmeasures this fractional part of a CKV cycle. As can be seen in FIG. 4b, for the steady state situation, without drift or modulation, theaddition of PHF and PHR will equal PHV and PHE will be zero.

FIG. 5 illustrates a block diagram of an ADPLL 80 designed to work offboth falling and rising edges of FREF. In this embodiment, the controlword FCW from FIG. 4a is halved, since it will be accumulated twice perFREF clock cycle (as shown by the retimed signal CKR×2, which is attwice the CKR frequency). The variable phase circuit 66 is also sampledtwice per FREF cycle. The fractional phase errors are computed using arising fractional phase circuit 64 a and a falling fractional phasecircuit 64 b. The rising fractional phase circuit determines a phaseerror on the rising edge of FREF and the falling fractional phasecircuit determines an intermediate phase error on the falling edge ofFREF. On the rising edge of FREF, the output of the rising fractionalphase error circuit 64 a is passed to adder 68 for the determination ofPHE, similar to that described in connection with FIGS. 4a-b . On thefalling edge of FREF, the intermediate phase error Φ₂′ is added to thefractional portion of the offset error ΔΦ (which could be greater thanone). The sum is then added to the integer portion of the offset errorΔΦ to provide Φ₂. This sum is passed to adder 68 for the computation ofPHE. Alternatively, ΔΦ could be added to Φ₂′, so long as the fractionalpart of ΔΦ is not compromised by a large value of ΔΦ.

FIG. 6 illustrates a time-to-digital converter (TDC) 90 for measuring afractional delay between CKV and FREF. Such a circuit may be used as therising or falling fractional phase circuits 64 a and 64 b. The CKVsignal passes through a string of inverters 92. Each inverter (or otherlogic device) 92 features a known time delay, for example, approximately20 psec using Texas Instruments' (CO35) of 0.08 □m L-effective CMOSprocess. On the active (rising) edge of FREF, a corresponding set ofregisters 94 capture the timing state output (D(1) . . . D(L)) of eachinverter. The output of every other register 94 is inverted tocompensate for the inversion by each inverter 92. The outputs (Q(1) . .. Q(L)) of the registers 94 are received by a pseudo-thermometer-codeedge detector 96, which outputs the location of a rising edge and afalling edge relative to FREF.

The operation of the circuit is shown in FIG. 7, which shows the Q(1 . .. L) values (L=10 in the illustrated embodiment) at the active edge ofFREF. In the illustrated example, the falling edge of CKV thatimmediately precedes FREF is displaced by two inverter delays, while therising edge of CKV that immediately precedes FREF is displaced by sixinverter delays.

While FIG. 5 shows two fractional phase detectors 64 a and 64 b, thefunctions of the two devices could be combined into a single circuitthat shares hardware, such as the string of inverters 92 and thepsuedo-thermometer code edge detector 96.

Additional detail on the operation of time-to-digital converter 90 canbe found in U.S. patent application Ser. No. 09/608,317, now U.S. Pat.No. 6,429,693, filed Jun. 30, 2000, entitled “Digital Fractional PhaseDetector” to Staszewski et al and in U.S. patent application Ser. No.09/967,275, now U.S. Pat. No. 6,593,773, filed Sep. 28, 2001, entitled“Power Saving Circuitry Using Predictive Logic” to Staszewski et al,both of which are incorporated by reference herein.

FIG. 8 illustrates a block diagram of a second embodiment of a PLL 100that can use both edges of a reference clock. In this embodiment, FREFis coupled to phase detector 102 (which is responsive to a first activeedge of FREF, e.g., the rising edge of FREF) and to phase detector 104(which is responsive to a second active edge of FREF, e.g., the fallingedge of FREF). The output of phase detector 102 is Φ1 and the output ofphase detector 104 is Φ₂. It is assumed that the phase detectors 102 and104 hold the output until the next compare event. The two most recentvalues of Φ1 and Φ2 are added by adder 106. The output of adder 106 isΦ_(E). Φ_(E) is passed to the loop filter 40 and to the controllableoscillator 42. The output of controllable oscillator 42 is fed back tothe phase detectors 102 and 104.

In operation, the controllable oscillator 42 updates the signal twiceper FREF clock cycle, driven by the average of the sum of the mostrecent outputs of the phase detectors as shown by Table 1. As in thecase of FIG. 4a , it is assumed that phase detector 102 compares therising edge of FREF with the rising edge of CKVD and phase detector 104compares the falling edge of FREF with the falling edge of CKVD. Itwould also be possible to compare the edges of FREF with the output CKV.In this case, both the rising edge and falling edge of FREF could becompared to a rising edge of CKV, if N (f_(CKV)/f_(FREF)) was an eveninteger. If N is an odd integer, rising edges of FREF could be comparedto rising edges of CKV and falling edges of FREF would be compared tofalling edges of CKV (it would also be possible to compare rising edgesof FREF to falling edges of CKV and vice-versa). Another possibilitywould b to add an offset Φ_(H), as described above.

TABLE 1 PHASE ERROR CALCULATION Clock cycle φ_(E) n (rising) φ₁(n) +φ₂(n − 1) n (falling) φ₁(n) + φ₂(n) n + 1 (rising) φ₁(n + 1) + φ₂(n) n +1 (falling) φ₁(n + 1) + φ₂(n + 1) n + 2 (rising) φ₁(n + 2) + φ₂(n + 1)n + 2 (falling) φ₁(n + 2) + φ₂(n + 2)

FIG. 9 shows a block diagram of a single sampling PLL 110 thatcalculates a phase error based on detections on both edges of FREF. Inthis embodiment, FREF is coupled to phase detector 112 and an invertedFREF is coupled to phase detector 114. The output of phase detector 112is Φ₁ and the output of phase detector 114 is Φ₂. The two most recentvalues of Φ₁ and Φ₂ are added by adder 116, but the total is latched bylatch 118 only on a single edge of FREF (the rising edge in theillustrated embodiment). The output of latch 118 is Φ_(E). Φ_(E) ispassed to the loop filter 40 and to the controllable oscillator 42. Theoutput of controllable oscillator 42 is fed back to the phase detectors112 and 114.

TABLE 2 PHASE ERROR CALCULATION Clock cycle φ_(E) n (rising) φ₁(n) +φ₂(n − 1) n + 1 (rising) φ₁(n + 1) + φ₂(n) n + 2 (rising) φ₁(n + 2) +φ₂(n + 1)

This embodiment differs from the embodiment of FIG. 8 in that the phaseerror driving the controllable oscillator 42 is updated only once perFREF cycle, although phase error contains components of updatesperformed twice during the FREF signal.

FIG. 10 illustrates a block diagram of a general mobile phonearchitecture 130. An RF transceiver 132 is coupled to an antenna 134 anda digital baseband circuit 136. The digital baseband circuit 136 iscoupled to a memory subsystem 138. In operation, the digital basebandcircuitry 136, which can include one or more digital signal processors(DSPs) and general purpose processors, generates the data fortransmission over the RF transceiver. As described above, the RFtransceiver uses a highly stable FREF signal, typically from a crystaloscillator. An external crystal 137 is coupled to a sync circuit 139(controlled in software by the digital baseband 136) that matches thefrequency and phase of the crystal (possibly through a PLL) with amaster clock (MCLK), which is embedded in communication data andbroadcast to the mobile device 130 by the base stations. In turn, thebase stations synchronize their master clock with an even more preciseclock signal, such as from a cesium clock, which may be received by thebase stations via fiber or satellite communications. The synchronizationcircuitry 139 can be very complex.

Additionally, there may be several clocks in the digital basebandcircuit 136 that run independently of the clocks in the RF transceiver132. This can cause significant noise, especially if the RF transceivercircuit were to be fabricated on the same circuit as the digitalbaseband circuit.

FIG. 11 illustrates a general block diagram of an architecture for theRF portion of a mobile communications device 140 which has significantadvantages over the prior art. A state machine 142 receives modulationand controllable oscillator timing update information (along withoptional feedback). State machine information is stored in latch 144,which is clocked at CKVD (a divided clock signal derived from the outputCKV). The output of latch 144 is coupled to the digitally controlledoscillator (DCO) 146. The output of DCO 146 is coupled to frequencydivider 148. The output of frequency divider 148 is CKVD.

In operation, the CKV signal is may be divided down by several frequencydividers 148 to provide suitable clock signals for devices in both theRF transceiver 132 and the digital baseband circuitry 136. For example,by generating a CKV having a frequency of 2.4 GHz, the signal could bedivided to a clock of about 8 MHz for generating data samples for aBluetooth applicaton and could be divided to a clock of about 100 MHzfor generating samples in an 802.11b application. Other divided clockfrequencies could be used for purposes other the data symbol generation.Preferably, the frequency dividers divide by a power of two.

FIG. 12 illustrates a more detailed block diagram of a first embodimentof the RF portion of the mobile communication device 140. In thisembodiment, symbols from the digital baseband circuitry 136 (see FIG.10) are received by data transmit modulation circuit 150. Data from thedata transmit modulation circuit 150 is output to state machine 142 andto digital-to-analog converter (DAC) 152, if amplitude modulation isbeing used. The output of DAC 152 drives power amplifier 154, which iscoupled to the output (CKV) of DCO 146. CKV is also output to frequencydivider 148 a, as well as to phase detector 156. The output of frequencydivider 148 a (CKVD1) is coupled to frequency dividers 148 b and 148 c,which output signals CKVD2 and CKVD3, respectively. CKVD2 clocks DAC 152and is also output to the data transmit modulation circuit 150 and todigital baseband circuitry 136. CKVD3 clocks latch 158 at the output ofphase detector 156. The output of latch 158 is coupled to state machine142.

In operation, the data modulation circuit creates sample points based onthe symbols received from the digital baseband circuit 136. In the priorart, a “chip clock” is used to generate these samples at a desiredfrequency. In general, the chip clock is a multiple of the referenceclock and requires a clock generation circuitry, such as a PLL togenerate a higher frequency clock from the reference clock. In theillustrated embodiment, however, a clock derived from the output of theDCO 146 (i.e., CKVD2) is used for the chip clock, by dividing the outputof the DCO. A data modulation circuit of this type is disclosed in U.S.patent application Ser. No. 10/001,448 to Staszewski et al, entitled“Transmit Filter”, filed Oct. 31, 2001, which is incorporated byreference herein. An apparent data rate can be adjusted by dynamicallychanging the oversampling ratio of the transmit filter.

Any number of clock frequencies could be generated from CKV. Theseclocks could be used in the various parts of a device, in particular inthe digital baseband circuit 136 and throughout the RF transceiver 132,eliminating multiple clock generation circuits. Additionally, usingclocks generated from CKV in both the baseband circuit 136 and in the RFtransceiver 132 provides many significant advantages. For example, sincethe clocks in the two subsystems are interrelated, operations occurringin the digital baseband circuit 136 could be timed to cause the leastpossible spurious noise in the RF transceiver 132.

FIG. 13 illustrates a second embodiment, similar to that of FIG. 12,where a single clock CKVD2 is used for both timing updates through thephase detector 156 and for data modulation through data transmitmodulation circuit 150. In this embodiment, Φ_(E) is generated by thephase detector 156 and passed to adder 160 such that a single number fordata modulation and timing updates is passed to state machine 142.

A value 4, an integer indicating a relative position of the CKVD2 clockto the free-running FREF frequency reference, is passed back to thedigital baseband circuit 136 where it may be used for synchronization,framing, timing adjustment of the fractional rate of the modulatingdata, and phase/frequency adjustment of the synthesized RF frequencywith MCLK. FREF can now operate in a free-running mode (i.e., notadjusted by MCLK). The master clock MCLK synchronization circuit wouldnow perform adjustment of the center frequency of the RF oscillator,rather than adjust FREF. Adjustments to the center frequency can be madeby adjustments to Φ_(E). Since FREF is a very stable clock and sinceMCLK updates are infrequent, adjustments are very small and occur overlong periods of time.

In FIGS. 12 and 13, the phase detection circuits are clocked by a signalthat is a division of CKV. Instead of using CKR, which is the FREF clockretimed to CKV clock (see FIG. 5), as the system clock, a power-of-twodivision (or any other division) of CKV could be used as the systemclock. Comparison events in the phase detector would be triggered by theFREF clock in which the clock timing delay is obtained bytime-to-digital converters (see FIG. 6) and the result used to performphase value adjustment.

As can be seen in FIGS. 12 and 13, some or all of the frequency dividers148 are controlled by a modulus control (MC) signal. This signal is usedto indicate a divisor. For example, for a divide-by-8/9 frequencydivider, MC=8/9. MC could vary dynamically during operation of thedevice in order to obtain a fractional division ratio.

Although the Detailed Description of the invention has been directed tocertain exemplary embodiments, various modifications of theseembodiments, as well as alternative embodiments, will be suggested tothose skilled in the art. The invention encompasses any modifications oralternative embodiments that fall within the scope of the Claims.

1. A radio frequency circuit comprising: a digitally controlledoscillator having a digital control input and an clock signal output;frequency divider circuitry having an input connected to the clocksignal output, a first divided clock output, a second divided clockoutput, and a third divided clock output; a latch having a latch input,a clock input coupled to the first divided clock signal output, and anoutput coupled to the digital control input; state machine circuitryhaving a modulation input, a timing update input, and a state output,the state output being coupled to the latch input; modulation circuitryhaving an input coupled to the second divided clock output and an outputcoupled to the modulation input; and timing update circuitry having aninput coupled to the clock signal output, the third divided clockoutput, and an output coupled to the timing update input.
 2. The radiofrequency circuit of claim 1 in which the frequency divider circuitryincludes: first divider circuitry having an input coupled with the clocksignal output and having the first divided clock output; second dividercircuitry having an input coupled with the first divided clock outputand having the second divided clock output; and third divider circuitryhaving an input coupled with the second divided clock output and havingthe third divided clock output.
 3. The radio frequency circuit of claim1 in which the second divided clock output is coupled to a digitalbaseband lead.
 4. The radio frequency circuit of claim 1 in which themodulation circuitry includes a symbol input receiving symbols fortransmission and a symbol output, and including a digital to analogconverter having a digital input coupled to the symbol output and ananalog output, and a power amplifier having a clock input coupled to theclock signal output, an analog input coupled to the analog output, and atransmission output.
 5. The radio frequency circuit of claim 1 in whichthe timing update circuitry includes a phase detection circuit having areference input coupled to a reference frequency lead, a clock inputcoupled to the clock signal output, and an output coupled to the timingupdate input.
 6. The radio frequency circuit of claim 1 in which thetiming update circuitry includes: a phase detection circuit having areference input coupled to a reference frequency lead, a clock inputcoupled to the clock signal output, and an output, and a latch having aninput coupled to the output of the phase detection circuit, a clockinput coupled to the third divided clock output, and an output coupledto the timing update input.